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PDF NCP1253 Data sheet ( Hoja de datos )
| Número de pieza | NCP1253 | |
| Descripción | Current-Mode PWM Controller | |
| Fabricantes | ON Semiconductor | |
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NCP1253
Current-Mode PWM
Controller for Off-line
Power Supplies
The NCP1253 is a highly integrated PWM controller capable of
delivering a rugged and high performance offline power supply in a
tiny TSOP−6 package. With a supply range up to 28 V, the controller
hosts a jittered 65 kHz or 100 kHz switching circuitry operated in peak
current mode control. When the power on the secondary side starts to
decrease, the controller automatically folds back its switching
frequency down to a minimum level of 26 kHz. As the power further
goes down, the part enters skip cycle while limiting the peak current.
To avoid sub harmonic oscillations in CCM operation, adjustable
slope compensation is available via the series inclusion of a simple
resistor in the current sense signal.
Besides the auto−recovery timer−based short−circuit protection, an
Over Voltage Protection on the VCC pin protects the whole circuitry in
case of optocoupler destruction or adverse open loop operation.
Features
• Fixed−Frequency 65 kHz or 100 kHz Current−Mode Control
Operation
• Frequency Foldback Down to 26 kHz and Skip−Cycle in Light Load
Conditions
• Adjustable Ramp Compensation
• Internally Fixed 4 ms soft−start
• Timer−based Auto−Recovery or Latched Short−Circuit Protection
• Frequency Jittering in Normal and Frequency Foldback Modes
• Latched OVP on VCC
• Up to 28 V VCC Operation
• Extremely Low No−load Standby Power
• These are Pb−Free Devices
Typical Applications
• Ac−dc Converters for TVs, Set−top Boxes and Printers
• Offline Adapters for Notebooks and Netbooks
www.onsemi.com
MARKING
DIAGRAM
TSOP−6
CASE 318G
1 STYLE 13
53xAYWG
G
1
53 = Specific Device Code
x = A, 2, C, or D
A = Assembly Location
Y = Year
W = Work Week
G = Pb−Free Package
(Note: Microdot may be in either location)
PIN CONNECTIONS
GND 1
6 DRV
FB 2
NC 3
5 VCC
4 CS
(Top View)
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 2 of this data sheet.
© Semiconductor Components Industries, LLC, 2014
December, 2014 − Rev. 1
1
Publication Order Number:
NCP1253/D
1 page  
NCP1253
ELECTRICAL CHARACTERISTICS
(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, Max TJ = 150°C, VCC = 12 V unless otherwise noted)
Symbol
Rating
Pin Min Typ Max
Unit
CURRENT COMPARATOR
Vfold
Vfreeze
TDEL
TLEB
TSS
Default internal voltage set point for frequency foldback trip point – 45% of
Vlimit
Internal peak current setpoint freeze (≈31% of Vlimit)
Propagation delay from current detection to gate off−state
Leading Edge Blanking Duration
Internal soft−start duration activated upon startup, auto−recovery
4
4
4
4
−
357 mV
250 mV
100 150
ns
300 ns
4 ms
INTERNAL OSCILLATOR
fOSC
Oscillation frequency (65 kHz version)
fOSC
Oscillation frequency (100 kHz version)
Dmax
Maximum duty−ratio
fjitter Frequency jittering in percentage of fOSC
fswing
Swing frequency
Feedback Section
− 61 65 71 kHz
− 92 100 108 kHz
− 76 80 84 %
− ±5
%
− 240 Hz
Rup Internal pull−up resistor
Req Equivalent ac resistor from FB to GND
Iratio Pin 2 to current setpoint division ratio
Vfreeze(FB) Feedback voltage below which the peak current is frozen
FREQUENCY FOLDBACK
2 20 kW
2 16 kW
− 4.2
2 1.05
V
Vfold
Frequency foldback level on the feedback pin – ≈45% of maximum peak
current
−
1.5
V
Ftrans
Vfold,end
Vskip
Skip
hysteresis
Transition frequency below which skip−cycle occurs
End of frequency foldback feedback level, Fsw = Fmin
Skip−cycle level voltage on the feedback pin
Hysteresis on the skip comparator
− 22 26 30 kHz
350 mV
− 300 mV
− 30 mV
INTERNAL SLOPE COMPENSATION
Vramp
Internal ramp level @ 25°C (Note 4)
4 2.5
V
Rramp Internal ramp resistance to CS pin
4 20 kW
4. A 1 MW resistor is connected from pin 4 to the ground for the measurement.
PROTECTIONS
VOVP
Latched Overvoltage Protection on the VCC rail
5 24 25.5 27
V
TOVPdel Delay before OVP confirmation on the VCC rail
5 20 ms
Timer Internal auto−recovery fault timer duration
− 100 130 160 ms
Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product
performance may not be indicated by the Electrical Characteristics if operated under different conditions.
www.onsemi.com
5
5 Page 
D2
1N4007
input
mains
D4
1N4007
NCP1253
R3
100k
D1
1N4007
D3
1N4007
Cbulk
22uF
Vcc
R2
100k
R1
200k
D6
1N4148
D5
1N4935
C1
4.7uF
C3
47uF
aux.
Figure 25. The Startup Resistor Can Be Connected to the Input Mains for Further Power Dissipation Reduction
The first step starts with the calculation of the needed VCC
capacitor which will supply the controller until the auxiliary
winding takes over. Experience shows that this time t1 can
be between 5 and 20 ms. Considering that we need at least
an energy reservoir for a t1 time of 10 ms, the Vcc capacitor
must be larger than:
CVCC
w
ICCt1
VCCon * VCCmin
w
3m
10m
w
3.3
(eq.
mF
1)
9
Let us select a 4.7 mF capacitor at first and experiments in
the laboratory will let us know if we were too optimistic for
t1. The VCC capacitor being known, we can now evaluate the
charging current we need to bring the Vcc voltage from 0 to
the VCCon of the IC, 18 V typical. This current has to be
selected to ensure a start−up at the lowest mains (85 V rms)
to be less than 3 s (2.5 s for design margin):
Icharge
w
VCCOnCVCC
2.5
w
18
4.7m
2.5
w
34
(eq.
mA
2)
If we account for the 15 mA that will flow inside the
controller, then the total charging current delivered by the
start−up resistor must be 49 mA. If we connect the start−up
network to the mains (half−wave connection then), we know
that the average current flowing into this start−up resistor
will be the smallest when VCC reaches the VCCon of the
controller:
If we account for the 15 mA that will flow inside the
controller, then the total charging current delivered by the
start−up resistor must be 49 mA. If we connect the start−up
network to the mains (half−wave connection then), we know
that the average current flowing into this start−up resistor
will be the smallest when VCC reaches the VCCon of the
controller:
ICVCC,min
+
Vac,rmsǸ2
p
*
VCCon
Rstart−up
(eq. 3)
To make sure this current is always greater than 49 mA, the
maximum value for Rstart−up can be extracted:
Rstart−up
v
Vac,rmsǸ2
p
*
VCCon
ICVCC,min
v
85
(eq. 4)
1.414
p
*
18
v
413
kW
49m
This calculation is purely theoretical, considering a
constant charging current. In reality, the take over time can
be shorter (or longer!) and it can lead to a reduction of the
Vcc capacitor. This brings a decrease in the charging current
and an increase of the start−up resistor, for the benefit of
standby power. Laboratory experiments on the prototype are
thus mandatory to fine tune the converter. If we chose the
400k resistor as suggested by Equation 4, the dissipated
power at high line amounts to:
ǒ ǓVac,peak 2
PRstart,max + 4Rstart−up +
320
4
Ǹ2 2
+
105k
400k 1.6Meg
(eq. 5)
+ 66 mW
Now that the first VCC capacitor has been selected, we
must ensure that the self−supply does not disappear when in
no−load conditions. In this mode, the skip−cycle can be so
deep that refreshing pulses are likely to be widely spaced,
inducing a large ripple on the VCC capacitor. If this ripple is
too large, chances exist to touch the VCCmin and reset the
controller into a new start−up sequence. A solution is to
grow this capacitor but it will obviously be detrimental to the
start−up time. The option offered in Figure 25 elegantly
www.onsemi.com
11
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| PDF Descargar | [ Datasheet NCP1253.PDF ] | |
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