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PDF SA5212 Data sheet ( Hoja de datos )
| Número de pieza | SA5212 | |
| Descripción | Transimpedance amplifier 140MHz | |
| Fabricantes | Philips | |
| Logotipo |  |
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INTEGRATED CIRCUITS
SA5212A
Transimpedance amplifier (140MHz)
Product specification
Replaces datasheet NE/SA/SE5212A of 1995 Apr 26
IC19 Data Handbook
1998 Oct 07
Philips
Semiconductors
1 page  
Philips Semiconductors
Transimpedance amplifier (140MHz)
Product specification
SA5212A
TEST CIRCUITS
SINGLE-ENDED
VOUT
Rt + VIN 2 @ S21 @ R
Ť Ť1 ) S22
RO + ZO 1 * S22 * 33
DIFFERENTIAL
VOUT
Rt + VIN 4 @ S21 @ R
Ť Ť1 ) S22
RO + 2ZO 1 * S22 * 66
SPECTRUM ANALYZER
NETWORK ANALYZER
S-PARAMETER TEST SET
PORT 1
PORT 2
ZO = 50Ω
VCC
0.1µF
OUT
R = 1k
IN DUT
OUT
1µF
33
33 1µF
50
GND1
GND2
RL = 50Ω
VCC
AV = 60DB
OUT
1µF
33
NC IN DUT
33 1µF
OUT
RL = 50
GND1
GND2
Test Circuit 1
Figure 2. Test Circuits 1 and 2
Test Circuit 2
SD00337
NETWORK ANALYZER
5V + ∆V
10µF
10µF
PORT 1
S-PARAMETER TEST SET
PORT 2
10µF
0.1µF
16
CURRENT PROBE
1mV/mA
VCC
33
OUT
NC IN DUT
33
OUT
GND1
GND2
1µF
1µF
CAL
100
BAL.
TRANSFORMER
NH0300HB
50 TEST
UNBAL.
Test Circuit 3
Figure 3. Test Circuit 3
SD00338
1998 Oct 07
5
5 Page 
Philips Semiconductors
Transimpedance amplifier (140MHz)
Product specification
SA5212A
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The SA5212A is a wide
bandwidth (typically 140MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The SA5212A is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 10. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
source to the emitter of Q3 is approximately the value of the
feedback resistor, RF=7kΩ. The gain from the second stage (A2)
and emitter followers (A3 and A4) is about two. Therefore, the
differential transresistance of the entire amplifier, RT is
RT
+
VOUT(diff)
IIN
+ 2RF
+ 2(7.2K)
+ 14.4kW
The single-ended transresistance of the amplifier is typically 7.2kΩ.
The simplified schematic in Figure 11 shows how an input current is
converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
the feedback resistor RF. The transistor Q1 provides most of the
open loop gain of the circuit, AVOL≈70. The emitter follower Q2
minimizes loading on Q1. The transistor Q4, resistor R7, and VB1
provide level shifting and interface with the Q15 – Q16 differential
pair of the second stage which is biased with an internal reference,
VB2. The differential outputs are derived from emitter followers Q11 –
Q12 which are biased by constant current sources. The collectors of
Q11 – Q12 are bonded to an external pin, VCC2, in order to reduce
the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 12, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
capacitance, CIN, in parallel with the source, IS, is approximately
7.5pF, assuming that CS=0 where CS is the external source
capacitance.
Since the input is driven by a current source the input must have a
low input resistance. The input resistance, RIN, is the ratio of the
incremental input voltage, VIN, to the corresponding input current, IIN
and can be calculated as:
RIN
+
VIN
IIN
+
1
RF
) AVOL
+
7.2K
70
+ 103W
More exact calculations would yield a higher value of 110Ω.
Thus CIN and RIN will form the dominant pole of the entire amplifier;
f*3dB
+
2p
1
RIN
CIN
Assuming typical values for RF = 7.2kΩ, RIN = 110Ω, CIN = 10pF
f*3dB
+
2p
(110)
1
10 @ 10*12
+ 145MHz
The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascade input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
a source capacitance of 1pF, input stage voltage gain of 70, RIN =
60Ω then the total input capacitance, CIN = (1+7.5) pF which will
lead to only a 12% bandwidth reduction.
OUTPUT +
A3
INPUT
A1
A2
RF A4
OUTPUT –
SD00327
Figure 10. SA5212A – Block Diagram
NOISE
Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of
3.5pA/√Hz. The transresistance configuration assures that the
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
equivalent input RMS noise current is strongly determined by the
quiescent current of Q1, the feedback resistor RF, and the
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 52nA RMS
in a 200MHz bandwidth.
1998 Oct 07
11
11 Page | ||
| Páginas | Total 20 Páginas | |
| PDF Descargar | [ Datasheet SA5212.PDF ] | |
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