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PDF NCV8871 Data sheet ( Hoja de datos )
| Número de pieza | NCV8871 | |
| Descripción | Automotive Grade Non-Synchronous Boost Controller | |
| Fabricantes | ON Semiconductor | |
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NCV8871
Automotive Grade
Non-Synchronous Boost
Controller
The NCV8871 is an adjustable output non−synchronous boost
controller which drives an external N−channel MOSFET. The device
uses peak current mode control with internal slope compensation. The
IC incorporates an internal regulator that supplies charge to the gate
driver.
Protection features include internally−set soft−start, undervoltage
lockout, cycle−by−cycle current limiting, hiccup−mode short−circuit
protection and thermal shutdown.
Additional features include low quiescent current sleep mode and
externally−synchronizable switching frequency.
Features
• Peak Current Mode Control with Internal Slope Compensation
• 1.2 V ±2% Reference voltage
• Fixed Frequency Operation
• Wide Input Voltage Range of 3.2 V to 40 Vdc, 45 V Load Dump
• Input Undervoltage Lockout (UVLO)
• Internal Soft−Start
• Low Quiescent Current in Sleep Mode
• Cycle−by−Cycle Current Limit Protection
• Hiccup−Mode Overcurrent Protection (OCP)
• Hiccup−Mode Short−Circuit Protection (SCP)
• Thermal Shutdown (TSD)
• This is a Pb−Free Device
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8
1
SOIC−8
D SUFFIX
CASE 751
MARKING
DIAGRAM
8
8871xx
ALYW
G
1
8871xx = Specific Device Code
xx = 00, 01, 02, 03, 04
A = Assembly Location
L = Wafer Lot
Y = Year
W = Work Week
G = Pb−Free Package
PIN CONNECTIONS
EN/SYNC 1
ISNS 2
GND 3
GDRV 4
8 VFB
7 VC
6 VIN
5 VDRV
(Top View)
ORDERING INFORMATION
Device
Package
NCV887100D1R2G SOIC−8
(Pb−Free)
Shipping†
2500 / Tape &
Reel
NCV887101D1R2G SOIC−8 2500 / Tape &
(Pb−Free)
Reel
NCV887102D1R2G SOIC−8 2500 / Tape &
(Pb−Free)
Reel
NCV887103D1R2G SOIC−8 2500 / Tape &
(Pb−Free)
Reel
NCV887104D1R2G SOIC−8 2500 / Tape &
(Pb−Free)
Reel
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
© Semiconductor Components Industries, LLC, 2016
August, 2016 − Rev. 12
1
Publication Order Number:
NCV8871/D
1 page  
NCV8871
ELECTRICAL CHARACTERISTICS (−40°C < TJ < 150°C, 3.2 V < VIN < 40 V, unless otherwise specified) Min/Max values are
guaranteed by test, design or statistical correlation.
Characteristic
Symbol
Conditions
Min Typ Max Unit
CURRENT SENSE AMPLIFIER
Current limit threshold voltage
Vcl
Voltage on ISNS pin
NCV887100
NCV887101
NCV887102
NCV887103
NCV887104
360 400 440
360 400 440
360 400 440
180 200 220
180 200 220
mV
Current limit,
Response time
Overcurrent protection,
Threshold voltage
tcl
%Vocp
CL tripped until GDRV falling edge,
VISNS = Vcl(typ) + 60 mV
Percent of Vcl
− 80 125
125 150 175
ns
%
Overcurrent protection,
Response Time
tocp From overcurrent event, Until switching
stops, VISNS = VOCP + 40 mV
VOLTAGE ERROR OPERATIONAL TRANSCONDUCTANCE AMPLIFIER
− − 125 ns
Transconductance
VEA output resistance
VFB input bias current
Reference voltage
VEA maximum output voltage
VEA minimum output voltage
VEA sourcing current
VEA sinking current
GATE DRIVER
gm,vea
Ro,vea
Ivfb,bias
Vref
Vc,max
Vc,min
Isrc,vea
Isnk,vea
VFB – Vref = ± 20 mV
Current out of VFB pin
VEA output current, Vc = 2.0 V
VEA output current, Vc = 0.7 V
0.8
2.0
−
1.176
2.5
−
80
80
1.2
−
0.5
1.200
−
−
100
100
1.63
−
2.0
1.224
−
0.3
−
−
mS
MW
mA
V
V
V
mA
mA
Sourcing current
Isrc VDRV ≥ 6 V, VDRV − VGDRV = 2 V
NCV887100
NCV887101
NCV887102
NCV887103
NCV887104
600 800
400 575
600 800
400 575
600 800
−
−
−
−
−
mA
Sinking current
Isink VGDRV ≥ 2 V
NCV887100
NCV887101
NCV887102
NCV887103
NCV887104
500 600
250 350
500 600
250 350
500 600
−
−
−
−
−
mA
Driving voltage dropout
Driving voltage source current
Backdrive diode voltage drop
Driving voltage
Vdrv,do
Idrv
Vd,bd
VDRV
VIN − VDRV, IvDRV = 25 mA
VIN − VDRV = 1 V
VDRV − VIN, Id,bd = 5 mA
NIVCDRVV88=701.010− 25 mA
NCV887101
NCV887102
NCV887103
NCV887104
− 0.3 0.6
35 45
−
− − 0.7
10 10.5 11
6.0 6.3 6.6
6.0 6.3 6.6
8.0 8.4 8.8
8.0 8.4 8.8
V
mA
V
V
UVLO
Undervoltage lock−out,
Threshold voltage
Vuvlo
VIN falling
3.0 3.1 3.2
V
Undervoltage lock−out,
Hysteresis
Vuvlo,hys
VIN rising
50 125 200 mV
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5
5 Page 
NCV8871
current at the maximum load at the worst case VIN, but
operation should be verified empirically. The worst case VIN
is half of VOUT, or whatever VIN is closest to half of VOUT.
After choosing a peak current ripple value, calculate the
inductor value as follows:
L
+
VIN(WC) DWC
DIL,max fs
Where: VIN(WC): VIN value as close as possible to
half of VOUT [V]
DWC: duty cycle at VIN(WC)
DIL,max: maximum peak to peak ripple [A]
The maximum average inductor current can be calculated
as follows:
IL,AVG
+
VOUTIOUT(max)
VIN(min)h
The Peak Inductor current can be calculated as follows:
IL,peak
+
IL,avg
)
DIL,max
2
Where: IL,peak: Peak inductor current value [A]
4. Select Output Capacitors
The output capacitors smooth the output voltage and
reduce the overshoot and undershoot associated with line
transients. The steady state output ripple associated with the
output capacitors can be calculated as follows:
VOUT(ripple) +
ǒ ǓDIOUT(max)
fCOUT
)
IOUT(max)
1*D
)
VIN(min)D
2fL
RESR
The capacitors need to survive an RMS ripple current as
follows:
Ǹ ǒ ǓICout(RMS) + IOUT
DWC
DȀWC
)
DWC
12
DȀWC
L
ROUT TSW
2
The use of parallel ceramic bypass capacitors is strongly
encouraged to help with the transient response.
5. Select Input Capacitors
The input capacitor reduces voltage ripple on the input to
the module associated with the ac component of the input
current.
ICin(RMS)
+
VIN(min) 2 DWC
LfsVOUT2 Ǹ3
6. Select Feedback Resistors
The feedback resistors form a resistor divider from the
output of the converter to ground, with a tap to the feedback
pin. During regulation, the divided voltage will equal Vref.
The lower feedback resistor can be chosen, and the upper
feedback resistor value is calculated as follows:
Rupper
+
Rlower
ǒVout * VrefǓ
Vref
The total feedback resistance (Rupper + Rlower) should be in
the range of 1 kW – 100 kW.
7. Select Compensator Components
Current Mode control method employed by the NCV8871
allows the use of a simple, Type II compensation to optimize
the dynamic response according to system requirements.
8. Select MOSFET(s)
In order to ensure the gate drive voltage does not drop out
the MOSFET(s) chosen must not violate the following
inequality:
Qg(total)
v
Idrv
fs
Where: Qg(total): Total Gate Charge of MOSFET(s) [C]
Idrv: Drive voltage current [A]
fs: Switching Frequency [Hz]
The maximum RMS Current can be calculated as follows:
IQ(max)
+
Iout
ǸD
DȀ
The maximum voltage across the MOSFET will be the
maximum output voltage, which is the higher of the
maximum input voltage and the regulated output voltaged:
VQ(max) + VOUT(max)
9. Select Diode
The output diode rectifies the output current. The average
current through diode will be equal to the output current:
ID(avg) + IOUT(max)
Additionally, the diode must block voltage equal to the
higher of the output voltage and the maximum input voltage:
VD(max) + VOUT(max)
The maximum power dissipation in the diode can be
calculated as follows:
PD + Vf (max) IOUT(max)
Where: Pd: Power dissipation in the diode [W]
Vf(max): Maximum forward voltage of the diode [V]
10. Determine Feedback Loop Compensation Network
The purpose of a compensation network is to stabilize the
dynamic response of the converter. By optimizing the
compensation network, stable regulation response is
achieved for input line and load transients.
Compensator design involves the placement of poles and
zeros in the closed loop transfer function. Losses from the
boost inductor, MOSFET, current sensing and boost diode
losses also influence the gain and compensation
expressions. The OTA has an ESD protection structure
(RESD ≈ 502 W, data not provided in the datasheet) located
on the die between the OTA output and the IC package
compensation pin (VC). The information from the OTA
PWM feedback control signal (VCTRL) may differ from the
IC-VC signal if R2 is of similar order of magnitude as RESD.
The compensation and gain expressions which follow take
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11
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| Páginas | Total 17 Páginas | |
| PDF Descargar | [ Datasheet NCV8871.PDF ] | |
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